Method and apparatus for sensing inductor input/output current in a dc-dc converter circuit

ABSTRACT

A switching regulator circuit has a switching transistor actuated during a switching on phase of a duty cycle. The current flowing through an inductor of the switching regulator circuit is determined from sensing a transistor current flowing through the switching transistor during switching on phase and selectively charging a capacitor of a switched capacitor circuit dependent on a current sense signal during the switching on phase.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation-in-part of U.S. patent application Ser. No. 18/225,912, filed Jul. 25, 2023, which is a continuation-in-part of U.S. patent application Ser. No. 17/748,214, filed May 19, 2022, the disclosures of which are incorporated herein by reference.

TECHNICAL FIELD

The present disclosure generally concerns switching regulators such as DC-DC converters and, in particular, the measurement of inductor (input or output) current in a switching regulator.

BACKGROUND

A switching regulator is a circuit that receives power at a first voltage level from a power source and outputs power at a second, different, voltage level for use by a downstream electronic system (the load). It is known in the art for a switching regulator to monitor input current in connection with carrying out various circuit operations and functions. For example, the switching regulator may include a current control loop that monitors the input current and uses the monitored input current as a control variable in a feedback loop to control the behavior of the switching regulator. It is also useful to be able to monitor the output current delivered to the load to determine, for example, the output power provided by the switching regulator.

In a buck-type converter, for example, one known solution for determining the output current of the switching regulator is to sense the currents flowing through the high-side (HS) transistor switch and low-side (LS) transistor switch. Output current can then be calculated by summing the high-side and low-side currents. Those skilled in the art recognize, however, that the measurement of the low-side current is adversely affected by switching dead time at the switching node (see, FIG. 1 which shows buck converter operation where the dead times are on the LS transistor switch), during which no current through the HS or LS transistor switches can be sensed, despite the fact that the real current contribution is non-zero, and thus the summation of the high-side and low-side currents is an inaccurate measure of the actual output current of the switching regulator. The dead times in the switching node introduce a critical error in the summation-based high-side and low-side current measurement process. This error increases with increases in the switching frequency of the switching regulator.

There is accordingly a need in the art for a method and apparatus that can more accurately determine the inductor (input or output) current of a switching regulator.

SUMMARY

In an embodiment, a switching regulator circuit comprises: a high side (HS) transistor coupled between a first node and a switching node; a low side (LS) transistor coupled between the switching node and a second node; driver circuitry for controlling actuation of the HS transistor and LS transistor in accordance with a duty cycle including a switching on phase and a switching off phase delimited by dead time; and an inductor having a first terminal coupled to the switching node and a second terminal coupled to a third node, wherein a coil current flows through the inductor.

A coil current detection circuit of the switching regulator circuit comprises: a current sensing circuit configured to sense a transistor current flowing through only one of the HS transistor and LS transistor during the switching on phase and generate a current sense signal; a first capacitor; a first switch configured to selectively charge the first capacitor in response to the current sense signal only during the switching on phase.

In an embodiment, a circuit for detecting a coil current flowing through an inductor of a switching regulator circuit having a switching transistor coupled to the inductor and driver circuitry for controlling actuation of the switching transistor in accordance with a duty cycle including a switching on phase and a switching off phase delimited by dead time comprises: a current sensing circuit configured to sense a transistor current flowing through the switching transistor during switching on phase of the duty cycle to generate a current sense signal indicative of the sensed transistor current; and a switched capacitor circuit actuated in response to the switching on phase to charge a capacitor dependent on the current sense signal

In an embodiment, a method for detecting a coil current flowing through an inductor of a switching regulator circuit having a switching transistor coupled to the inductor and driver circuitry for controlling actuation of the switching transistor in accordance with a duty cycle including a switching on phase and a switching off phase delimited by dead time comprises: sensing a transistor current flowing through the switching transistor during switching on phase to generate a current sense signal indicative of the sensed transistor current; and selectively charging a capacitor of a switched capacitor circuit dependent on the current sense signal during the switching on phase.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing and other features and advantages will be discussed in detail in the following non-limiting description of specific embodiments in connection with the accompanying drawings, in which:

FIG. 1 shows a timing diagram for the voltage at the switching node of a buck-type switching regulator;

FIG. 2 is a block diagram showing a system including a buck-type switching regulator and an output current detection circuit that operates to determine the output current of the switching regulator;

FIG. 3 illustrates waveforms for currents and voltages at certain nodes of the switching regulator shown in FIG. 2 ;

FIG. 4 is a circuit diagram for a current sensing circuit of the output current detection circuit in FIG. 2 ;

FIGS. 5A and 5B are circuit diagram for embodiments of an amplifier used in a current sensing circuit;

FIGS. 6 and 7 are circuit diagrams for a duty cycle detect and divide circuit of the output current detection circuit in FIG. 2 ;

FIG. 8 is a block diagram showing a system including a boost-type switching regulator and an input current detection circuit that operates to determine the input current of the switching regulator;

FIG. 9 illustrates waveforms for currents and voltages at certain nodes of the switching regulator shown in FIG. 8 ;

FIG. 10 is a circuit diagram for a current sensing circuit of the input current detection circuit in FIG. 8 ;

FIGS. 11 and 12 are circuit diagrams for a duty cycle detect and divide circuit of the input current detection circuit in FIG. 8 ;

FIG. 13 is a block diagram showing a system including a buck-type switching regulator and an output current detection circuit that operates to determine the output current of the switching regulator;

FIG. 14 illustrates waveforms for input and coil currents for a buck-type switching regulator operating in continuous conduction mode (CCM);

FIG. 15 is a circuit diagram for a current sensing circuit of the output current detection circuit in FIG. 13 used in CCM;

FIG. 16 illustrates waveforms for input and coil currents for a buck-type switching regulator operating in discontinuous conduction mode (DCM);

FIG. 17 is a circuit diagram for a current sensing circuit of the output current detection circuit in FIG. 13 used in DCM;

FIG. 18 is a block diagram showing a system including a boost-type switching regulator and an input current detection circuit that operates to determine the input current of the switching regulator;

FIG. 19 is a circuit diagram for a current sensing circuit of the input current detection circuit in FIG. 18 used in CCM; and

FIG. 20 is a circuit diagram for a current sensing circuit of the input current detection circuit in FIG. 18 used in DCM.

DETAILED DESCRIPTION

The same elements have been designated with the same reference numerals in the different drawings. In particular, the structural and/or functional elements common to the different embodiments may be designated with the same reference numerals and may have identical structural, dimensional, and material properties.

Throughout the present disclosure, the term “connected” is used to designate a direct electrical connection between circuit elements with no intermediate elements other than conductors, whereas the term “coupled” is used to designate an electrical connection between circuit elements that may be direct, or may be via one or more intermediate elements.

The terms “about”, “substantially”, and “approximately” are used herein to designate a tolerance of plus or minus 10%, preferably of plus or minus 5%, of the value in question.

The determination of the output current of a buck-type switching regulator can advantageously be made dependent on a measurement of the current I_(AVG) (HS) flowing through the high-side (HS) transistor switch during the switching on phase (T_(ON)). The average current in the inductor of the switching regulator (i.e., the regulator output current) is then extrapolated solely from the sensed high-side transistor switch current using the switching duty cycle set by the ratio of the switching on phase (T_(ON)) to the switching off phase (T_(OFF)) for the HS transistor switch.

Consider, for example, the following for a buck-type switching regulator. The average current I_(AVG) (LS) flowing through the low-side (LS) transistor switch is given by:

${I_{AVG}\left( {LS} \right)} = {{I_{AVG}\left( {HS} \right)}*\frac{T_{OFF}}{T_{ON}}}$

The output current I_(OUT) of the switching regulator is then:

I _(OUT) =I _(AVG)(HS)+I _(AVG)(LS)

Substituting gives:

$I_{OUT} = {{{I_{AVG}\left( {HS} \right)}*\left( {1 + \frac{T_{OFF}}{T_{ON}}} \right)} = {{I_{AVG}\left( {HS} \right)}*\left( \frac{T_{ON} + T_{OFF}}{T_{ON}} \right)}}$

Where the real duty cycle D for switching the high-side (HS) transistor switch and low-side (LS) transistor switch is:

$D = \frac{T_{ON}}{T_{ON} + T_{OFF}}$

In this context, the “real” duty cycle means the duty cycle defined by the real times when the HS transistor switch is on (T_(ON)) and when the LS transistor switch is on (Torr), not the ideal duty cycle (V_(OUT)/V_(IN)).

Simplified:

$I_{OUT} = \frac{I_{AVG}\left( {HS} \right)}{D}$

Thus, by sensing the current I_(AVG) (HS) flowing through the high-side (HS) transistor switch, and with knowledge of the real duty cycle D of the switching regulator, the output current of the switching regulator can be determined.

The current I_(AVG)(HS) can be measured through a power-power sense circuit (described in more detail below) and the real duty cycle D can be extrapolated from the ratio T_(OFF)/T_(ON) through the drive signals which control switching of the high-side (HS) transistor and low-side (LS) transistor switches. A division of the measured current I_(AVG) (HS) by the real duty cycle D of the switching regulator is then made by a detect and divide circuit (described in more detail below) to generate a value corresponding to the inductor output current of the switching regulator.

Reference is now made to FIG. 2 which shows a block diagram for an output current detection circuit 100 that operates to determine the output current of a switching regulator 102 (shown here to be of the buck-type, by example), and further to FIG. 3 which shows waveforms (in continuous conduction mode (CCM) operation) for currents and voltages at certain nodes of the buck-type switching regulator shown in FIG. 2 . The switching regulator 102 includes a high-side (HS) transistor switch 104 (for example, implemented as a p-channel MOS transistor) having a source terminal coupled, preferably directly connected, to an input voltage Vin and a drain terminal coupled, preferably directly connected, to a switching node 106. The gate of the HS transistor switch 104 is driven by a high side driver circuit 108 that outputs a high side control signal 110 (Vgate H s). A current I_(HS) flows through the switch 104. The switching regulator 102 further includes a low-side (LS) transistor switch 114 (for example, implemented as an n-channel MOS transistor) having a drain terminal coupled, preferably directly connected, to the switching node 106 and a source terminal coupled, preferably directly connected, to a reference voltage node (for example, ground). The gate of the LS transistor switch 114 is driven by a low side driver circuit 118 that outputs a low side control signal 120 (Vgate_(LS)). A current I_(LS) flows through the switch 114. In a preferred embodiment, the high side control signal 110 and the low side control signal 120 have a same frequency and are non-overlapping pulse width modulated (PWM) signals where the high side control signal 110 controls the HS transistor switch 104 to turn on during a switching on phase (T_(ON)) and turn off during a switching off phase (T_(OFF)). In an embodiment, the low side control signal 120 may be derived from the high side control signal 110 with a small margin (referred to as dead time 122) to avoid both transistors being on at the same time. The low side control signal 120 controls the LS transistor switch 114 to turn on during a portion of the switching off phase (T_(OFF)) which is delimited by the dead time 122 due to the non-overlapping switch control. An inductor 130 has a first terminal coupled, preferably directly connected, to the switching node 106 and a second terminal coupled, preferably directly connected, to an output node. A current/coil flows through the inductor 130. A load circuit (represented here by a load capacitance and resistance in parallel) is coupled to the output node.

The output current detection circuit 100 includes a high side current sensing circuit 140 configured to sense the current I_(AVG) (HS), also referred to as the current I_(HS), flowing through the HS transistor switch 104 and output a signal indicative of the sensed current I_(AVG) (HS) sense A duty cycle detect and divide circuit (1/D) 180 receives a high side control signal 110 b and a low side control signal 120 b. The signals 110 b and 120 b are derived from (or related to) the signals 110 and 120, respectively, which control the switching on phase (T_(ON)) and switching off phase (T_(OFF)) but have a shorter dead time 122 b than the dead time 122 for the signals 110 and 120 (see, FIG. 2 ) and are thus more accurately representative of the real duty cycle D of the switching regulator operation. The signal indicative of the sensed current I_(AVG)(HS)_(sense) is then divided by the determined real duty cycle D to generate an output signal that is indicative of the extrapolated value I_(OUTextrap) for the output current I_(OUT) of the switching regulator.

Reference is now made to FIG. 4 which shows a circuit diagram for an embodiment of the current sensing circuit 140. A first input transistor M1 (for example, implemented as a p-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the source terminal (s) of the HS transistor switch 104 (at the input voltage Vin) and a drain terminal coupled, preferably directly connected, to a first (inverting (−)) input of a differential amplifier 142. The gate terminal of the first input transistor M1 is driven by the high side control signal 110. A second input transistor M2 (for example, implemented as a p-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the drain terminal (d) of the HS transistor switch 104 (at the switching node 106) and a drain terminal coupled, preferably directly connected, to a second (non-inverting (+)) input of the differential amplifier 142. The gate terminal of the second input transistor M2 is also driven by the high side control signal 110. Input transistors M1 and M2 are controlled by the high side control signal 110 to turn on during the switching on phase (T_(ON)). An output transistor M3 (for example, implemented as a p-channel MOS transistor) has a source terminal coupled, preferably directly connected, in a negative feedback circuit to the first (inverting) input of the differential amplifier 142. A resistor 144 with a fixed resistance R has a first terminal coupled, preferably directly connected, to a drain terminal of the output transistor M3 and a second terminal coupled, preferably directly connected, to the reference voltage node (for example, ground). A passive low-pass filter circuit 148 formed by the series circuit connection of a resistor 150 and capacitor 152 is coupled, preferably directly connected, between the drain terminal of the output transistor M3 and the reference voltage node. A series connection node 156 of the low-pass filter circuit 148 between the resistor 150 and capacitor 152 provides an output of the current sensing circuit 140 producing the signal Vsense indicative of the sensed current I_(AVG)(HS)_(sense). The first and second input transistors M1, M2 function to sample the source and drain voltages, respectively, of the HS transistor switch 104 responsive to the high side control signal 110 only during switching on phase (T_(ON)) for application to the differential amplifier 142. The difference between those sampled voltages is converted by the differential amplifier 142 and output transistor M3 to a current I_(sense) which replicates the current flowing through the inductor 130 during switching on phase (T_(ON)). This current I_(sense) is converted to a voltage by resistor 144 and the voltage is smoothed by the low pass filter circuit 148 to generate the output voltage signal Vsense.

FIG. 5A shows a circuit diagram for an embodiment of the differential amplifier 142. A first amplifier transistor M4 (for example, implemented as a p-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the inverting (−) input of the differential amplifier 142 and a drain terminal biased by a first current sink circuit 160 configured to sink a current I_(bias) to the reference voltage node. A second amplifier transistor M5 (for example, implemented as a p-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the non-inverting (+) input of the differential amplifier 142 and a drain terminal biased by a second current sink circuit 162 configured to sink the current I_(bias) to the reference voltage node. The gate terminals of the transistors M4 and M5 are coupled, preferably directly connected, to each other and to the drain terminal of the transistor M4.

Let K be a ratio of power to power sense. During the switching on phase (T_(ON)) when the high side control signal 110 is asserted logic low, transistors 104, M1 and M2 are all turned on. If the gain of the differential amplifier 142 is high enough to ensure that the voltages at the inverting (−) input and non-inverting (+) input of the differential amplifier 142 are substantially equal (through the negative feedback with transistor M3), then the following is true:

V ₁₀₆ −K*R _(on) *I _(bias) =Vin−K*R _(on)*(I _(bias) +I _(sense))

Where: V₁₀₆ is the voltage at the switching node 106, R on is drain-to-source resistance of the transistor 104 in the ON state (i.e., Rds_on), I_(bias) is the bias current for the current sink circuits 160, 162, and I_(sense) is the current flowing through the transistor M3.

Then, the sensed current I_(AVG) (HS)I_(sense) is:

${I_{AVG}\left( {HS} \right)}_{sense} = {\frac{{V{in}} - V_{106}}{K*R_{on}} = {\frac{I_{AVG}\left( {HS} \right)}{K} = \frac{I_{out\_ on}}{K}}}$

Where: I_(out_on) is the average current I_(AVG)(HS) flowing through the inductor 130 during the time period T_(ON)+T_(OFF)+T_(HiZ).

It will accordingly be noted that the current I_(out_on) and its replica current I_(sense) (i.e., I_(AVG) (HS)_(sense)) increase with a ratio of K. In this context, K is substantially constant, and it allows to have the current I_(sense) be directly proportional to the current flowing through the inductor 130 when the HS transistor switch 104 is in the ON state. It is important that the value for K is large enough to have the magnitude of I_(sense) in a range of some micro-amperes, despite an output current I_(out_on) that has a magnitude on the order of amperes.

The signal at the output of the current sensing circuit 140 that is indicative of the sensed current I_(AVG)(HS)_(sense) is actually a low pass filtered voltage Vsense that is equal to R * I_(AVG) (HS) sense where R is the resistance of the resistor 144. Such low pass filter is effective when its time constant is substantially longer than the total switching time R₁₅₀ C₁₅₂>>T_(SW)=T_(ON)+T_(OFF)+T_(HiZ).

Reference is now made to FIG. 6 which shows a circuit diagram for an embodiment of the duty cycle detect and divide circuit 180. A differential amplifier 182 includes a non-inverting (+) input coupled to receive the voltage Vsense (which is indicative of the sensed current I_(AVG)(HS)_(sense) in the HS transistor switch 104 during the switching on phase (T_(ON))). The output of the differential amplifier 182 is coupled, preferably directly connected, to the inverting (−) input in negative feedback by a capacitor 184. This forms a unity gain analog integration circuit that functions to integrate (i.e., average) the difference in voltages at the non-inverting (+) input and inverting (−) input. Such integration is effective when the time constant is substantially longer than the total switching time: R₁ C₁₈₄>>T_(SW). The voltage at the non-inverting (+) input is the voltage Vsense. The voltage at the inverting (−) input is a switched voltage selected by a voltage switching circuit 185 controlled by the ratio

$\frac{T_{ON}}{T_{OFF}}$

to be equal to either the voltage V_(current_out) at the output of the differential amplifier 182 or the reference voltage (i.e., ground voltage).

The voltage switching circuit 185 includes a first switching transistor M6 (for example, implemented as a p-channel MOS transistor) having a source terminal coupled, preferably directly connected, to the output of the differential amplifier 182 and a drain terminal coupled, preferably directly connected, to a switching node 186. The first switching transistor M6 is controlled by signal 110 b to turn on during the T_(ON) state (i.e., when transistor 104 is turned on by the high side control signal 110). Thus, the gate of transistor M6 may be configured to receive the control signal 110 b and is turned on when control signal 110 b is asserted (logic low). A second switching transistor M7 (for example, implemented as an n-channel MOS transistor) has a drain terminal coupled, preferably directly connected, to the switching node 186 and a source terminal coupled, preferably directly connected, to the reference voltage node. The second switching transistor M7 is controlled by signal 120 b to turn on during the T_(OFF) state (i.e., when transistor 104 is turned off by the high side control signal 110). Thus, the gate of transistor M7 may be configured to receive the control signal 120 b and is turned on when control signal 120 b is asserted (logic high).

The signal generated at the switching node 186 is equal, on average, to D * V_(current_out), where V_(current_out) is the voltage at the output of the differential amplifier 182. When the first switching transistor M6 is turned on during the switching on phase (T_(ON)), the voltage V_(current_out) at the output of the differential amplifier 182 is applied to the inverting (−) input of the differential amplifier 182. The differential amplifier 182 will respond to this by driving the voltage V_(current_out) toward the voltage V_(sense)−V_(current_out) Conversely, when the second switching transistor M7 is turned on during the switching off phase (T_(OFF)), the reference voltage (ground) is instead applied to the inverting (−) input of the differential amplifier 182. The differential amplifier 182 will respond to this by driving the voltage V_(current_out) toward the voltage Vsense. Over time, due to the averaging function provided by the unity gain integration on capacitor 184, the voltage V_(current_out) at the output of the differential amplifier 182 will settle at a value equal to:

$V_{current\_ out} = {{V{sense}*\left( \frac{T_{ON} + T_{OFF}}{T_{ON}} \right)} = \frac{V{sense}}{D}}$

In other words, supposing that the voltage at the (+) and (−) inputs of the amplifier 182 are equal, the following equation applies:

$V_{current\_ out} = {{V{sense}*\frac{1}{D}} = {{\frac{R}{K}\frac{1}{D}I_{HS}} = {\frac{R}{K}I_{OUT}}}}$

Where: R is the resistance of resistor 144 of FIG. 4 .

The voltage V_(current_out) at the output of the differential amplifier 182 is the signal indicative of the extrapolated value I_(OUTextrap) for the output current I_(OUT) of the switching regulator.

Reference is now made to FIG. 7 which shows a circuit diagram for another embodiment of the duty cycle detect and divide circuit 180. A differential amplifier 192 includes a non-inverting (+) input coupled to receive the voltage Vsense (which is indicative of the sensed current I_(AVG)(HS)_(sense) in the HS transistor switch 104 during the switching on phase (T_(ON))). The output of the differential amplifier 192 is coupled, preferably directly connected, to the inverting (−) input in negative feedback by a resistor 194 having a fixed resistance Rb. The inverting (−) input is further coupled to ground through a resistor 196 having a variable resistance Ra. The value of the variable resistance Ra is digitally controlled by the bits of a digital selection signal 198 generated by an analog-to-digital converter (ADC) circuit 200. This circuit is essentially a non-inverting operational amplifier with a controllable gain dependent on the resistances Ra and Rb (where the amplifier gain

$\left. {G = \left( {1 + \frac{Rb}{Ra}} \right)} \right).$

A voltage switching circuit 202 includes a first switching transistor M8 (for example, implemented as a p-channel MOS transistor) having a source terminal coupled, preferably directly connected, to a supply voltage Vdd and a drain terminal coupled, preferably directly connected, to a switching node 204. The first switching transistor M8 is controlled by signal 110 b to turn on during the T_(ON) state (i.e., when transistor 104 is turned on by the high side control signal 110). Thus, the gate of transistor M8 may be configured to receive the control signal 110 b and is turned on when control signal 110 b is asserted (logic low). A second switching transistor M9 (for example, implemented as an n-channel MOS transistor) has a drain terminal coupled, preferably directly connected, to the switching node 204 and a source terminal coupled, preferably directly connected, to the reference voltage node (ground). The second switching transistor M9 is controlled by signal 120 b to turn on during the T_(OFF) state (i.e., when transistor 104 is turned off by the high side control signal 110). Thus, the gate of transistor M9 may be configured to receive the control signal 120 b and is turned on when control signal 120 b is asserted (logic high). The switching node 204 is coupled to the input of the ADC circuit 200 through a low-pass filter circuit formed by resistor 206 and capacitor 208.

The voltage at the switching node 204 is a switched voltage selected by the voltage switching circuit 202 controlled by the ratio

$\frac{T_{ON}}{T_{OFF}}$

to be equal to either the supply voltage Vdd or the reference voltage (i.e., ground voltage). Analog filtering of the switched voltage by the low-pass filter circuit produces an input voltage to the ADC circuit 200 equal, on average, to D*Vdd. This switched and filtered voltage is converted by the ADC circuit 200 to generate the digital selection signal 198 for controlling the variable resistance Ra of the resistor 196. The resistances Ra, Rb of the resistors 196, 194 set the gain of the differential amplifier 192 applied to the voltage Vsense (indicative of the sensed current I_(AVG)(HS)_(sense) in the HS transistor switch 104 during the switching on phase (T_(ON))) which is applied to the non-inverting (+) input.

The following relationship is achieved through the non-inverting amplifier circuit using the differential amplifier 192 with resistors 194 and 196:

$V_{current\_ out} = {V{sense}\left( {1 + \frac{Rb}{Ra}} \right)}$

The variable resistance Ra of the resistor 196 is digitally controlled to satisfy the following relationship:

${1 + \frac{Rb}{Ra}} = \frac{1}{D}$

Because Ra must satisfy the foregoing equation, this means that:

${Ra} = {\frac{D}{1 - D}Rb}$

Where the resistance Rb is fixed. If we consider, as an example, the ADC circuit 200 with a three-bit resolution, the input signal D*Vdd is then quantized to eight levels, with the following Table defining the relationship of the variable resistance Ra as a function of the duty cycle and the fixed resistance Rb:

D Ra 0 0 1/8 Rb/7 2/8 Rb/3 3/8 3*Rb/5 4/8 Rb 5/8 5*Rb/3 6/8 3*Rb 7/8 7*Rb

The output of the ADC circuit 200 controls the variable resistor 196 to select the correct resistance Ra. This can be accomplished, for example, using an analog multiplexer having mux inputs connected to resistances for the possible values of Ra, a control input connected to the output of the ADC circuit 200 and a mux output connected to resistor Rb and the input of amplifier 192.

It will be noted that the implementation with a three-bit resolution is just an example, and this concept is extendible to a higher resolution to provide higher accuracy in the selection of the variable resistance value.

Because of the foregoing relationship between D, Ra and Rb, the following equation applies:

$V_{current\_ out} = {{V{sense}*\frac{1}{D}} = {{\frac{R}{K}\frac{1}{D}I_{HS}} = {\frac{R}{K}I_{OUT}}}}$

Where: R is the resistance of resistor 144 in FIG. 4 .

The voltage V_(current_out) at the output of the differential amplifier 192 is the signal indicative of the extrapolated value I_(OUTextrap) for the output current I_(OUT) of the switching regulator.

FIG. 3 illustrates operation of the buck converter in continuous conduction mode (CCM). It will, however, be noted that the current sensing circuit 140 and duty cycle detect and divide circuit 180 can also operate to sense inductor output current during ON time when the converter is instead configured for operation in discontinuous conduction mode (DCM). In this case, beyond the ON and OFF times, a high-impedance state (High-Z) is present at the switching node when the coil current is zero. In this high-impedance state: transistors 104, 114 are turned off, transistors M6, M7 are turned off, and transistors M8, M9 are turned off.

The foregoing technique for buck-type regulator circuit output current determination is also applicable in the context of determining the input current of a boost-type regulator circuit.

The determination of the input current of a boost-type switching regulator can advantageously be made dependent on a measurement of the current I_(AVG)(LS) flowing through the low-side (LS) transistor switch during the switching on phase (T_(ON)). The average current in the inductor of the switching regulator (i.e., the regulator input current) is then extrapolated solely from the sensed low-side current using the switching duty cycle set by the ratio of the switching on phase (T_(ON)) to the switching off phase (Torr) for the LS transistor switch.

Consider, for example, the following for a boost-type switching regulator. The average current I_(AVG) (HS) flowing through the high-side (HS) transistor switch is given by:

${I_{AVG}\left( {HS} \right)} = {{I_{AVG}\left( {LS} \right)}*\frac{T_{OFF}}{T_{ON}}}$

The input current I_(IN) of the switching regulator is then:

I _(IN) =I _(AVG)(HS)+I _(AVG)(LS)

Substituting gives:

$I_{IN} = {{{I_{AVG}\left( {LS} \right)}*\left( {1 + \frac{T_{OFF}}{T_{ON}}} \right)} = {{I_{AVG}\left( {LS} \right)}*\left( \frac{T_{ON} + T_{OFF}}{T_{ON}} \right)}}$

Where the real duty cycle D for switching the high-side (HS) transistor and low-side (LS) transistor switches is:

$D = \frac{T_{ON}}{T_{ON} + T_{OFF}}$

In this context, the “real” duty cycle means the duty cycle defined by the real times when the LS transistor switch is on (T_(ON)) and when the HS transistor switch is on (T_(OFF)), not the ideal duty cycle (V_(OUT)/V_(IN)).

Simplified:

$I_{IN} = \frac{I_{AVG}\left( {LS} \right)}{D}$

Thus, by sensing the current I_(AVG)(LS) flowing through the low-side (LS) transistor switch, and with knowledge of the real duty cycle D of the switching regulator, the input current of the switching regulator can be determined.

The current I_(AVG) (LS) can be measured through a power-power sense circuit (described in more detail below) and the real duty cycle D can be extrapolated from the ratio T_(ON)/T_(OFF) through the drive signals which control switching of the low-side (LS) transistor and high-side (HS) transistor switches. A division of the measured current I_(AVG) (LS) by the real duty cycle D of the switching regulator is then made by a detect and divide circuit (described in more detail below) to generate a value corresponding to the input current of the switching regulator.

Reference is now made to FIG. 8 which shows a block diagram for an input current detection circuit 300 that operates to determine the input current of a switching regulator 302 (shown here to be of the boost-type, by example), and further to FIG. 9 which shows waveforms (in continuous conduction mode (CCM) operation) for currents and voltages at certain nodes of the switching regulator shown in FIG. 8 . The switching regulator 302 includes a high-side (HS) transistor switch 304 (for example, implemented as a p-channel MOS transistor) having a source terminal coupled, preferably directly connected, to an output node and a drain terminal coupled, preferably directly connected, to a switching node 306. The gate of the HS transistor switch 304 is driven by a high side driver circuit 308 that outputs a high side control signal 310 (Vgate_(HS)). A current I_(HS) (also referred to the as the regulator output current I_(OUT)) flows through the switch 304 to an output node. A load circuit (represented here by a load capacitance and resistance in parallel) is coupled to the output node. The switching regulator 302 further includes a low-side (LS) transistor switch 314 (for example, implemented as an n-channel MOS transistor) having a drain terminal coupled, preferably directly connected, to the switching node 306 and a source terminal coupled, preferably directly connected, to a reference voltage node (for example, ground). The gate of the LS transistor switch 314 is driven by a low side driver circuit 318 that outputs a low side control signal 320 (Vgate_(LS)). A current I_(LS) flows through the switch 314. In a preferred embodiment, the high side control signal 310 and the low side control signal 320 have a same frequency and are non-overlapping pulse width modulated (PWM) signals where the low side control signal 320 controls the LS transistor switch 314 to turn on during a switching on phase (T_(ON)) and turn off during a switching off phase (T_(OFF)). In an embodiment, the high side control signal 310 may be derived from the low side control signal 320 with a small margin (referred to as dead time) to avoid both transistors being on at the same time. The high side control signal 310 controls the HS transistor switch 304 to turn on during a portion of the switching off phase (Tori) which is delimited by a dead time 322 due to the non-overlapping switch control. An inductor 330 has a first terminal coupled, preferably directly connected, to the switching node 306 and a second terminal coupled, preferably directly connected, to receive an input voltage Vin. A current I_(coil) (also referred to herein as an input current T_(IN)) flows through the inductor 330.

The input current detection circuit 300 includes a low side current sensing circuit 340 configured to sense the current I_(AVG)(LS), also referred to as the current I_(LS), flowing through the LS transistor switch 314 and output a signal indicative of the sensed current I_(AVG)(LS)_(sense) A duty cycle detect and divide circuit (1/D) 380 receives a high side control signal 310 b and a low side control signal 320 b. The signals 310 b and 320 b are derived from (related to) the signals 310 and 320, respectively, which control the switching on phase (T_(ON)) and switching off phase (Torr) but have a shorter dead time 322 b than the dead time 322 for the signals 310 and 320 (see, FIG. 8 ) and are thus more accurately representative of the real duty cycle D of the switching regulator operation. The signal indicative of the sensed current I_(AVG) (LS) sense is then divided by the determined real duty cycle D to generate an output signal that is indicative of the extrapolated value I_(INextrap) for the input current I_(IN) of the switching regulator.

Reference is now made to FIG. 10 which shows a circuit diagram for an embodiment of the current sensing circuit 340. A first input transistor M11 (for example, implemented as an n-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the drain terminal (d) of the LS transistor switch 314 (at the switching node 306) and a drain terminal coupled, preferably directly connected, to a first (non-inverting (+)) input of a differential amplifier 342. The gate terminal of the first input transistor M11 is driven by the low side control signal 320. A second input transistor M12 (for example, implemented as an n-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the source terminal (s) of the LS transistor switch 314 (at the reference node; ground) and a drain terminal coupled, preferably directly connected, to a second (inverting (−)) input of the differential amplifier 342. The gate terminal of the second input transistor M12 is also driven by the low side control signal 320. Input transistors M11 and M12 are controlled by the low side control signal 320 to turn on during the switching on phase (T_(ON)). An output transistor M13 (for example, implemented as an n-channel MOS transistor) has a source terminal coupled, preferably directly connected, in a negative feedback circuit to the second (inverting) input of the differential amplifier 342. A drain terminal of the output transistor M13 is connected to an input leg of a current mirroring circuit 343 (formed by mirror connected p-channel MOS transistors). A resistor 344 has a first terminal coupled, preferably directly connected, to an output leg of the current mirroring circuit 343 and a second terminal coupled, preferably directly connected, to the reference voltage node (for example, ground). A passive low-pass filter circuit 348 formed by the series circuit connection of a resistor 350 and capacitor 352 is coupled, preferably directly connected, between the output leg of the current mirroring circuit 343 and the reference voltage node. A series connection node 356 of the low-pass filter circuit 348 between the resistor 350 and capacitor 352 provides an output of the current sensing circuit 340 producing the signal indicative of the sensed current I_(AVG)(LS)_(sense) The first and second input transistors M11, M12 function to sample the drain and source voltages, respectively, of the LS transistor switch 314 responsive to the low side control signal 320 only during switching on phase (T_(ON)) for application to the differential amplifier 342. The difference between those sampled voltages is converted by the differential amplifier 342 and output transistor M13 to a current I_(sense) which replicates the current flowing through the inductor 330 during switching on phase (T_(ON)). This current I_(sense) is mirrored by the current mirroring circuit 343 and converted to a voltage by resistor 344 and the voltage is smoothed by the low pass filter circuit 348 to generate an output voltage Vsense.

FIG. 5B shows a circuit diagram for an embodiment of the differential amplifier 342. A first amplifier transistor M14 (for example, implemented as an n-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the inverting (−) input of the differential amplifier 342 and a drain terminal biased by a first current source circuit 161 configured to source a current I_(bias) from a supply voltage node. A second amplifier transistor M15 (for example, implemented as an n-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the non-inverting (+) input of the differential amplifier 342 and a drain terminal biased by a second current source circuit 163 configured to source the current I_(bias) from the supply voltage node. The gate terminals of the transistors M14 and M15 are coupled, preferably directly connected, to each other and to the drain terminal of the transistor M14.

Let K be a ratio of power to power sense. During the switching on phase (T_(ON)) when the low side control signal 320 is asserted, transistors 314, M11 and M12 are all turned on. If the gain of the differential amplifier 342 is high enough to ensure that the voltages at the inverting (−) input and non-inverting (+) input of the differential amplifier 342 are substantially equal (through the negative feedback with transistor M13), the following is true:

V ₃₀₆ −K*R _(on) *I _(bias) −K*R _(on)*(I _(bias) −I _(sense))

Where: V 306 is the voltage at the switching node 306, R on is drain-to-source resistance of the transistor 314 in the ON state (i.e., Rds_on), I_(bias) is the bias current for the current source circuits 161, 163, and I_(sense) is the current flowing through the transistor M13.

Then, the sense current I_(sense) is:

$I_{sense} = {\frac{V_{306}}{K*R_{on}} = \frac{I_{LS}}{K}}$

Where: I_(LS) is the current flowing through the inductor 330 when the transistor 314 in the ON state.

It will accordingly be noted that the current I_(LS) and its replica current I_(sense) increase with a ratio of K. In this context, K is substantially constant, and it allows to have the current I_(sense) directly proportional to the current flowing through the inductor 330 when the LS transistor switch 314 is in the ON state. It is important that the value for K is large enough to have the magnitude of I_(sense) in a range of some micro-amperes, despite an inductor current I_(LS) that has a magnitude on the order of amperes.

After mirroring by the current mirroring circuit 343, the signal at the output of the current sensing circuit 340 that is indicative of the sensed current I_(AVG)(L S)_(sense) is actually a low pass filtered voltage Vsense that is equal to R * I_(AVG)(LS)_(sense) where R is the resistance of the resistor 344. Such low pass filter is effective when its time constant is substantially longer than the total switching time R₃₅₀ C₃₅₂>>T_(SW)=T_(ON)+T_(OFF)+T_(HIZ).

Reference is now made to FIG. 11 which shows a circuit diagram for an embodiment of the duty cycle detect and divide circuit 380. A differential amplifier 382 includes a non-inverting (+) input coupled to receive the voltage Vsense (which is indicative of the sensed current I_(AVG)(LS)_(sense) in the LS transistor switch 314 during the time period T_(ON)+T_(OFF)+T_(HIZ)). The output of the differential amplifier 382 is coupled, preferably directly connected, to the inverting (−) input in negative feedback by a capacitor 384. This forms a unity gain analog integration circuit that functions to integrate (i.e., average) the difference in voltages at the non-inverting (+) input and inverting (−) input. The voltage at the non-inverting (+) input is the voltage Vsense. The voltage at the inverting (−) input is a switched voltage selected by a voltage switching circuit 385 controlled by the ratio

$\frac{T_{ON}}{T_{OFF}}$

to be equal to either the voltage V_(current_out) at the output of the differential amplifier 382 or the reference voltage (i.e., ground voltage).

The voltage switching circuit 385 includes a first switching transistor M16 (for example, implemented as a p-channel MOS transistor) having a source terminal coupled, preferably directly connected, to the output of the differential amplifier 382 and a drain terminal coupled, preferably directly connected, to a switching node 386. The first switching transistor M16 is controlled by a logical inverse of the signal 320 b to turn on during T_(ON) state (i.e., when transistor 314 is turned on by the low side control signal 320). Thus, the gate of transistor M16 may be configured to receive the logical inverse of the signal 320 b and is turned on when control signal 320 b is asserted (logic high). A second switching transistor M17 (for example, implemented as an n-channel MOS transistor) has a drain terminal coupled, preferably directly connected, to the switching node 386 and a source terminal coupled, preferably directly connected, to the reference voltage node. The second switching transistor M17 is controlled by a logical inverse of the signal 310 b during T_(OFF) state (i.e., when transistor 314 is turned off by the low side control signal 320). Thus, the gate of transistor M17 may be configured to receive the logical inverse of signal 310 b and is turned on when control signal 310 b is asserted (logic low). The switching node 386 is coupled in negative feedback through resistor R1 to the inverting (−) input of the differential amplifier 382. The signal generated at the switching node 386 is equal, on average, to D * V_(current_out), where V_(current_out) is the voltage at the output of the differential amplifier 382. When the first switching transistor M16 is turned on during the switching on phase (T_(ON)), the voltage V_(current_out) at the output of the differential amplifier 382 is applied to the inverting (−) input of the differential amplifier 382. The differential amplifier 382 will respond to this by driving the voltage V_(current_out) toward the voltage V_(sense)−V_(current_out) Conversely, when the second switching transistor M17 is turned on during the switching off phase (T_(OFF)), the reference voltage (ground) is instead applied to the inverting (−) input of the differential amplifier 382. The differential amplifier 382 will respond to this by driving the voltage V_(current_out) toward the voltage Vsense. Over time, due to the averaging function provided by the unity gain integration on capacitor 384, the voltage V_(current_out) at the output of the differential amplifier 382 will settle at a value equal to:

$V_{current\_ out} = {{V{sense}*\left( \frac{T_{ON} + T_{OFF}}{T_{ON}} \right)} = \frac{V{sense}}{D}}$

In other words, supposing that the voltage at the (+) and (−) inputs of the amplifier 382 are equal, the following equation applies:

$V_{current\_ out} = {{V{sense}*\frac{1}{D}} = {{\frac{R}{K}\frac{1}{D}I_{LS}} = {\frac{R}{K}I_{IN}}}}$

Where: R is the resistance of resistor 344 in FIG. 10 .

The voltage V_(current_out) at the output of the differential amplifier 382 is the signal indicative of the extrapolated value I_(INextrap) for the input current I_(IN) of the switching regulator.

Reference is now made to FIG. 12 which shows a circuit diagram for another embodiment of the duty cycle detect and divide circuit 380. A differential amplifier 392 includes a non-inverting (+) input coupled to receive the voltage Vsense (which is indicative of the sensed current I_(AVG)(LS)_(sense) in the LS transistor switch 314 during the time period T_(ON)+T_(OFF)+T_(HIZ)). The output of the differential amplifier 392 is coupled, preferably directly connected, to the inverting (−) input in negative feedback by a resistor 394 having a fixed resistance Rb. The inverting (−) input is further coupled to ground through a resistor 396 having a variable resistance Ra. The value of the variable resistance Ra is digitally controlled by the bits of a digital selection signal 398 generated by an analog-to-digital converter (ADC) circuit 400. This circuit is essentially a non-inverting operational amplifier with a controllable gain dependent on the resistances Ra and Rb (where the amplifier gain

$\left. {G = \left( {1 + \frac{Rb}{Ra}} \right)} \right).$

A voltage switching circuit 401 includes a first switching transistor M18 (for example, implemented as a p-channel MOS transistor) having a source terminal coupled, preferably directly connected, to a supply voltage Vdd and a drain terminal coupled, preferably directly connected, to a switching node 304. The first switching transistor M18 is controlled by a logical inverse of signal 320 b to turn on during T_(ON) state (i.e., when transistor 314 is turned on by the low side control signal 320). Thus, the gate of transistor M18 may be configured to receive the logical inverse of signal 320 b and is turned on when control signal 320 b is asserted (logic high). A second switching transistor M19 (for example, implemented as an n-channel MOS transistor) has a drain terminal coupled, preferably directly connected, to the switching node 304 and a source terminal coupled, preferably directly connected, to the reference voltage node (ground). The second switching transistor M19 is controlled by a logical inverse of signal 310 b to turn on during T_(OFF) state (i.e., when transistor 314 is turned off by the low side control signal 320). Thus, the gate of transistor M19 may be configured to receive the logical inverse of signal 310 b and is turned on when control signal 310 b is asserted (logic low). The switching node 304 is coupled to the input of the ADC circuit 400 through a low-pass filter circuit formed by resistor 306 and capacitor 308.

The voltage at the switching node 404 is a switched voltage selected by the voltage switching circuit 401 controlled by the ratio

$\frac{T_{ON}}{T_{OFF}}$

to be equal to either the supply voltage Vdd or the reference voltage (i.e., ground voltage). Analog filtering of the switched voltage by the low-pass filter circuit produces an input voltage to the ADC circuit 400 equal, on average, to D*Vdd. This switched and filtered voltage is converted by the ADC circuit 400 to generate the digital selection signal 398 for controlling the variable resistance Ra of the resistor 396. The resistances Ra, Rb of the resistors 396, 394 set the gain of the differential amplifier 392 applied to the voltage Vsense (indicative of the sensed current I_(AVG)(LS)_(sense) in the LS transistor switch 314 during the time period T_(ON)+T_(OFF)+T_(HIZ)) which is applied to the non-inverting (+) input.

The following relationship is achieved through the non-inverting amplifier circuit using the differential amplifier 392 with resistors 394 and 396:

$V_{current\_ out} = {V{sense}\left( {1 + \frac{Rb}{Ra}} \right)}$

The variable resistance Ra of the resistor 396 is digitally controlled to satisfy the following relationship:

${1 + \frac{Rb}{Ra}} = \frac{1}{D}$

Because Ra must satisfy the foregoing equation, this means that:

${Ra} = {\frac{D}{1 - D}Rb}$

Where the resistance Rb is fixed. If we consider, as an example, the ADC circuit 400 with a three-bit resolution, the input signal D*Vdd is then quantized to eight levels, with the following Table defining the relationship of the variable resistance Ra as a function of the duty cycle and the fixed resistance Rb:

D Ra 0 0 1/8 Rb/7 2/8 Rb/3 3/8 3*Rb/5 4/8 Rb 5/8 5*Rb/3 6/8 3*Rb 7/8 7*Rb

The output of the ADC circuit 400 controls the variable resistor 396 to select the correct resistance Ra. This can be accomplished, for example, using an analog multiplexer having inputs connected to resistances for the possible values of Ra, a control input connected to the output of the ADC circuit 400 and an output connected to resistor Rb and the input of amplifier 392.

It will be noted that the implementation with a three-bit resolution is just an example, and this concept is extendible to a higher resolution to provide higher accuracy in the selection of the variable resistance value.

Because of the foregoing relationship between D, Ra and Rb, the following equation applies:

$V_{current\_ out} = {{V{sense}*\frac{1}{D}} = {{\frac{R}{K}\frac{1}{D}I_{LS}} = {\frac{R}{K}I_{IN}}}}$

Where: R is the resistance of resistor 344 of FIG. 10 .

The voltage V_(current_out) at the output of the differential amplifier 392 is the signal indicative of the extrapolated value I_(INextrap) for the input current I_(IN) of the switching regulator.

Reference is once again made to FIG. 8 . The output current I_(OUT) of the switching regulator is the current that flows through the high side transistor 304 to the load. This current is the difference between the input current I_(IN) of the switching regulator and the current I_(LS) of the low side transistor 314. To extract a value for the output current I_(OUT), and generate a signal indicative of the extrapolated value I_(OUTextrap) for the output current I_(OUT) of the switching regulator, a summation circuit subtracts the sensed current I_(AVG)(LS) sense in the LS transistor switch 314 from the extrapolated value I_(INextrap) for the input current I_(IN).

FIG. 9 illustrates operation of the boost converter in continuous conduction mode (CCM). It will, however, be noted that the current sensing circuit 340 and duty cycle detect and divide circuit 380 can also operate for sensing inductor input current during ON time where the converter is instead configured for operation in discontinuous conduction mode (DCM). In this case, beyond the ON and OFF times, a high-impedance state is present when the coil current is zero. In this high-impedance state: transistors 304, 314 are turned off, transistors M16, M17 are turned off, and transistors M18, M19 are turned off.

Although illustrated herein by example only for the calculation of switching regulator output current for a buck-type circuit (FIG. 2 ) and the calculation of switching regulator input current for a boost-type circuit (FIG. 8 ), it will be understood that the technique disclosure herein is equally applicable to current detection in other types of switching regulator circuits (buck-boost, inverting, etc.).

Reference is now made to FIG. 13 which shows a block diagram for an output current detection circuit 403 that operates to determine the output current of a switching regulator 402 (shown here to be of the buck-type, by example), and further to FIG. 14 which shows waveforms (in continuous conduction mode (CCM) operation) for the input current his and coil current ‘OUT of the buck-type switching regulator shown in FIG. 13 . The switching regulator 402 includes a high-side (HS) transistor switch 404 (for example, implemented as a p-channel MOS transistor) having a source terminal coupled, preferably directly connected, to an input voltage Vin and a drain terminal coupled, preferably directly connected, to a switching node 406. The gate of the HS transistor switch 404 is driven by a high side driver circuit 408 that outputs a high side control signal 410. A current I_(HS) flows through the switch 404. The switching regulator 402 further includes a low-side (LS) transistor switch 414 (for example, implemented as an n-channel MOS transistor) having a drain terminal coupled, preferably directly connected, to the switching node 406 and a source terminal coupled, preferably directly connected, to a reference voltage node (for example, ground). The gate of the LS transistor switch 414 is driven by a low side driver circuit 418 that outputs a low side control signal 420. A current I_(LS) flows through the switch 414. In a preferred embodiment, the high side control signal 410 and the low side control signal 420 have a same frequency and are non-overlapping pulse width modulated (PWM) signals where the high side control signal 410 controls the HS transistor switch 404 to turn on during a switching on phase (T_(ON)) and turn off during a switching off phase (T OFF). In an embodiment, the low side control signal 420 may be derived from the high side control signal 410 with a small margin (referred to as dead time 422) to avoid both transistors being on at the same time. The low side control signal 420 controls the LS transistor switch 414 to turn on during a portion of the switching off phase (T OFF) which is delimited by the dead time 422 due to the non-overlapping switch control. An inductor 430 has a first terminal coupled, preferably directly connected, to the switching node 406 and a second terminal coupled, preferably directly connected, to an output node. A current I_(OUT) a flows through the inductor 430. A load circuit (represented here by a load capacitance and resistance in parallel) is coupled to the output node.

With reference to FIG. 14 , it will be noted that for the buck regulator shown in FIG. 13 operating in continuous conduction mode (CCM), the coil current I_(OUT) is triangularly shaped, and the average of the input current I_(HS) flowing through the high-side (HS) transistor switch 104 during T_(ON) state is equal to the average of the coil current I_(OUT) during the switching period for CCM (this average current value being identified in FIG. 14 as I_(avg)). Because of this equivalence, the coil current I_(OUT) can be determined from detection of the input current Lis.

The output current detection circuit 403 includes a coil current detection circuit 440 with a high side current sensing circuit configured to sense the current I_(HS) flowing through the HS transistor switch 404 and output a signal that is indicative of the extrapolated value I_(OUTextrap) for the output current I_(OUT) of the switching regulator.

Reference is now made to FIG. 15 which shows a circuit diagram for an embodiment of the coil current detection circuit 440. A first input transistor M21 (for example, implemented as a p-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the source terminal (s) of the HS transistor switch 404 (at the input voltage Vin) and a drain terminal coupled, preferably directly connected, to a first (inverting (−)) input of a differential amplifier 442. The gate terminal of the first input transistor M21 is driven by the high side control signal 410. A second input transistor M22 (for example, implemented as a p-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the drain terminal (d) of the HS transistor switch 404 (at the switching node 406) and a drain terminal coupled, preferably directly connected, to a second (non-inverting (+)) input of the differential amplifier 442. The gate terminal of the second input transistor M22 is also driven by the high side control signal 410. Input transistors M21 and M22 are controlled by the high side control signal 410 to turn on during the switching on phase (T_(ON)). An output transistor M23 (for example, implemented as a p-channel MOS transistor) has a source terminal coupled, preferably directly connected, in a negative feedback circuit to the first (inverting) input of the differential amplifier 442. The foregoing circuitry forms a high side current sensing circuit. A resistor 444 with a fixed resistance R has a first terminal coupled, preferably directly connected, to a drain terminal of the output transistor M23 and a second terminal coupled, preferably directly connected, to the reference voltage node (for example, ground). A switch transistor M24 (for example, implemented as an n-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the drain terminal of the output transistor M23 and a drain terminal coupled, preferably directly connected, to an output 456 of the coil current detection circuit 440 producing the signal Vsense indicative of the output current Iota of the switching regulator. The gate terminal of the switch transistor M24 is driven by a signal derived from the logical inversion of the high side control signal 410. A capacitor 452 is coupled, preferably directly connected, between the output 456 (at the drain terminal of the switch transistor M24) and the reference voltage node. The capacitor 452 is selectively charged in response to the current I_(sense) when the switch transistor M24 is turned on by the logical inversion of the high side control signal 410 during the switching on phase (T_(ON)). The charging current for capacitor 452 is equal to the current I_(sense) minus the current flowing through the resistor 444.

The first and second input transistors M21, M22 function to sample the source and drain voltages, respectively, of the HS transistor switch 404 responsive to the high side control signal 410 only during switching on phase (T_(ON)) for application to the differential amplifier 442. The difference between those sampled voltages is converted by the differential amplifier 442 and output transistor M23 to a current I_(sense) which replicates the current flowing through the inductor 430 during switching on phase (T_(ON)). This current I_(sense) is converted to a voltage by resistor 444. That voltage is selectively filtered in response to actuation of the switch transistor M24 using capacitor 452 to generate the output voltage signal Vsense.

An advantage of the circuit implementation shown in FIGS. 13 and 15 is that it obviates the need for the duty cycle detect and divide circuit 180 used in the circuit implementation of FIGS. 2, 4, 6 and 7 .

FIG. 5A shows a circuit diagram for an embodiment of the differential amplifier 442.

Let K be a ratio of power to power sense. During the switching on phase (T_(ON)) when the high side control signal 410 is asserted logic low, transistors 404, M21, M22 and M24 are all turned on. If the gain of the differential amplifier 442 is high enough to ensure that the voltages at the inverting (−) input and non-inverting (+) input of the differential amplifier 142 are substantially equal (through the negative feedback with transistor M23), then the following is true:

V ₄₀₆ −K*R _(on) *I _(bias) =Vin−K*R _(on)*(I _(bias) −I _(sense))

Where: V 406 is the voltage at the switching node 406, R on is drain-to-source resistance of the transistor 404 in the ON state (i.e., Rds_on), I_(bias) is the bias current for the current sink circuits 160, 162, and I_(sense) is the current flowing through the transistor M23.

In this context, K is substantially constant, and it allows to have the current I_(sense) be directly proportional to the current flowing through the inductor 430 when the HS transistor switch 404 is in the ON state. It is important that the value for K is large enough to have the magnitude of I_(sense) in a range of some micro-amperes, despite an output current I_(OUT) that has a magnitude on the order of amperes.

Due to the switching operation of the switch transistor M24, filtering is performed by capacitor 452 only during switching on phase (T_(ON)). The average of the current in the capacitor 452 is zero at steady state. Considering the capacitance of the capacitor 452 to be high enough that the voltage across the capacitor (Vsense) is substantially constant, the current I_(c) flowing through the capacitor 452 during switching on phase (T_(ON)) is given by the following formula:

${I_{c}(t)} = {{I_{sense}(t)} - \frac{V{sense}}{R}}$

Where: R is the resistance of the resistor 444.

Then:

${\frac{1}{T_{ON}}{\int_{0}^{T_{ON}}{{I_{sense}(t)}dt}}} = \frac{V{sense}}{R}$

The capacitance C 452 must be selected large enough such that R C₄₅₂>>T_(ON) because the equation above is valid only if Vsense is approximately constant during the ON time.

Solving for Vsense:

${Vsense} = {{\frac{R}{K}\frac{1}{T_{ON}}{\int_{0}^{T_{ON}}{{I_{HS}(t)}dt}}} = {\frac{R}{K}I_{OUT}}}$

From the foregoing, the output voltage Vsense from circuit 440, which is indicative of the output current I_(OUT), is directly proportional to the average of the current flowing through the HS transistor switch 404 responsive to assertion of the high side control signal 410 during switching on phase (T_(ON)).

Reference is now made to FIG. 16 . For the buck regulator shown in FIG. 13 operating in discontinuous conduction mode (DCM), the coil current I_(OUT) is triangularly shaped, and the average (I_(avg-HS)) of the input current Ills flowing through the high-side (HS) transistor switch 404 during T_(ON) state is not equal to the average (I_(avg_OUT)) of the coil current I_(OUT) during the switching period for DCM. The reason for this is that the high-impedance (High-Z) state of controlling switching (where both of the transistors 404 and 414 are turned off and a high impedance condition exists at the switching node 406) introduces a contribution to the average coil current in which the coil current is zero. The circuit solution shown in FIG. 15 for the coil current detection circuit 440 would not produce an accurate output when operating in DCM.

Reference is now made to FIG. 17 which shows a circuit diagram for a modified embodiment of the coil current detection circuit 440 that is used in connection with DCM operation. A first input transistor M21 (for example, implemented as a p-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the source terminal (s) of the HS transistor switch 404 (at the input voltage Vin) and a drain terminal coupled, preferably directly connected, to a first (inverting (−)) input of a differential amplifier 442. The gate terminal of the first input transistor M21 is driven by the high side control signal 410. A second input transistor M22 (for example, implemented as a p-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the drain terminal (d) of the HS transistor switch 404 (at the switching node 406) and a drain terminal coupled, preferably directly connected, to a second (non-inverting (+)) input of the differential amplifier 442. The gate terminal of the second input transistor M22 is also driven by the high side control signal 410. Input transistors M21 and M22 are controlled by the high side control signal 410 to turn on during the switching on phase (T_(ON)). An output transistor M23 (for example, implemented as a p-channel MOS transistor) has a source terminal coupled, preferably directly connected, in a negative feedback circuit to the first (inverting) input of the differential amplifier 442. The foregoing circuitry forms a high side current sensing circuit. A resistor 444 with a fixed resistance R has a first terminal coupled, preferably directly connected, to a drain terminal of the output transistor M23 and a second terminal coupled, preferably directly connected, to the reference voltage node (for example, ground). A first switch transistor M24 (for example, implemented as an n-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the drain terminal of the output transistor M23 and a drain terminal coupled, preferably directly connected, to a first intermediate node 470. The gate terminal of the first switch transistor M24 is driven by a signal derived from a logical inversion of the high side control signal 410. A capacitor 452 is coupled, preferably directly connected, between node 470 (at the drain terminal of the first switch transistor M24) and the reference voltage node. The capacitor 452 is selectively charged in response to the current I_(sense) when the switch transistor M24 is turned on by the logical inversion of the high side control signal 410 during the switching on phase (T_(ON)). The charging current for capacitor 452 is equal to the current I_(sense) minus the current flowing through the resistor 444. A second switch transistor M25 (for example, implemented as an n-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the node 470 (at the drain terminal of the first switch transistor M24) and a drain terminal coupled, preferably directly connected, to a second intermediate node 472. The gate terminal of the second switch transistor M25 is driven by a signal derived from the logical inversion of a control signal High-Z, generated by a timing control circuit, which is logic high when the regulator circuit is in the high-impedance (High-Z) state and is logic low when the regulator circuit is not in the high-impedance (High-Z) state. A third switch transistor M26 (for example, implemented as an n-channel MOS transistor) has a drain terminal coupled, preferably directly connected, to the node 472 (at the drain terminal of the second switch transistor M25) and a source terminal coupled, preferably directly connected, to the reference voltage node. The gate terminal of the third switch transistor M26 is driven by a control signal High-Z′, generated by the timing control circuit, which is derived from the signal High-Z but with a small dead time (for example, on the order of a few hundred pico-seconds in order to avoid cross-conduction with transistors M25 and M26). A resistor 474 is coupled between the node 472 and the output 456 of the coil current detection circuit 440 which produces the signal Vsense indicative of the output current Iota of the switching regulator. A capacitor 476 is coupled, preferably directly connected, between the output 456 and the reference voltage node. The resistor 474 and capacitor 476 form a passive low-pass filter circuit. Such low pass filter is effective if its time constant is sufficiently larger than the total switching period R₄₇₄ C₄₇₆>>T_(SW).

The circuitry formed by the first switching transistor M24 and capacitor 452 operates, analogous to that discussed above in connection with FIG. 15 , to generate a signal at node 470 that is proportional to an average of the input current I_(HS) flowing through the high side transistor 404 during switching on phase (T_(ON)).

The circuitry formed by the second and third switching transistors M25, M26 functions as a masking circuit that masks (or blocks) the contribution of the signal at node 470 during the high impedance (High-Z) state. During the high impedance (High-Z) state, the switching transistor M25 is deactuated (blocking the signal at node 470 from passing to node 472) and the switching transistor M26 is actuated (connecting node 472 to ground). Conversely, when not in the high impedance (High-Z) state, the switching transistor M25 is actuated (passing the signal at node 470 to node 472 to be filtered by resistor 474 and capacitor 476) and the switching transistor M26 is deactuated (disconnecting node 472 from ground).

The circuitry formed by resistor 474 and capacitor 476 filters the signal at node 472 to generate the output voltage Vsense which is proportional to the output current I_(OUT) of the switching regulator.

Operationally, the average of the currents flowing through capacitors 452 and 476 must be zero. Thus, the following system of equations applies:

$\left\{ \begin{matrix} {{{\int_{0}^{T_{ON}}{{I_{sense}(t)}dt}} - {\frac{V_{A}}{R_{444}}T_{ON}} + {\frac{{Vsense} - V_{A}}{R_{474}}\left( {T_{ON} + T_{OFF}} \right)}} = 0} \\ {{{\frac{V_{A} - {Vsense}}{R_{474}}\left( {T_{ON} + T_{OFF}} \right)} - {\frac{Vsense}{R_{474}}\left( {T_{SW} - T_{ON} - T_{OFF}} \right)}} = 0} \end{matrix} \right.$

Where: V_(A) is the voltage at node 470; R₄₄₄ is the resistance of resistor 444; R₄₇₄ is the resistance of resistor 474; T_(ON) is the on time; Torr is the off time; and T_(SW) is the overall switching period which includes T_(ON), T_(OFF) and High-Z (where High-Z is the time where neither of the transistors 404 or 414 is actuated).

Assuming that either the resistance R474 is very high, or that there is voltage buffering using voltage follower 480, this simplifies to:

$\left\{ \begin{matrix} {{{\int_{0}^{T_{ON}}{{I_{sense}(t)}dt}} - {\frac{V_{A}}{R_{444}}T_{ON}}} = 0} \\ {{{\frac{V_{A} - {Vsense}}{R_{474}}\left( {T_{ON} + T_{OFF}} \right)} - {\frac{Vsense}{R_{474}}\left( {T_{SW} - T_{ON} - T_{OFF}} \right)}} = 0} \end{matrix} \right.$

Solving this system of equations for Vsense:

${Vsense} = {\frac{\int_{0}^{T_{ON}}{{I_{HS}(t)}{dt}}}{K\left\lbrack {\frac{1}{R_{444}}T_{SW}\frac{T_{ON}}{T_{ON} + T_{OFF}}} \right\rbrack} = {\frac{R_{444}}{K}\frac{1}{T_{SW}}\frac{T_{ON} + T_{OFF}}{T_{ON}}{\int_{0}^{T_{ON}}{{I_{HS}(t)}dt}}}}$

Where:

$I_{OUT} = {\frac{T_{ON} + T_{OFF}}{T_{SW}}\frac{1}{T_{ON}}{\int_{0}^{T_{ON}}{{I_{HS}(t)}dt}}}$

Substituting:

${Vsense} = {\frac{R_{444}}{K}I_{OUT}}$

Thus, the voltage of the output signal Vsense is directly proportional to the output current I_(OUT) flowing in the inductor.

As noted above, the solution here is based on the assumption that the resistance R₄₇₄ for resistor 474 is substantially high or a unity gain buffer circuit 480 is inserted in the signal path between nodes 470 and 472 in series with the second switch transistor M25.

The foregoing technique for buck-type regulator circuit output current determination is also applicable in the context of determining the input current of a boost-type regulator circuit.

The determination of the input current of a boost-type switching regulator can advantageously be made dependent on a measurement of the current I_(LS) flowing through the low-side (LS) transistor switch during the switching on phase (T_(ON)). The average current in the inductor of the switching regulator (i.e., the regulator input current) is then extrapolated solely from the sensed low-side current.

Reference is now made to FIG. 18 which shows a block diagram for an input current detection circuit 500 that operates to determine the input current of a switching regulator 502 (shown here to be of the boost-type, by example) configured for operation in continuous conduction mode (CCM). The switching regulator 502 includes a high-side (HS) transistor switch 504 (for example, implemented as a p-channel MOS transistor) having a source terminal coupled, preferably directly connected, to an output node and a drain terminal coupled, preferably directly connected, to a switching node 506. The gate of the HS transistor switch 504 is driven by a high side driver circuit 508 that outputs a high side control signal 510. A current I_(HS) (also referred to as the regulator output current I_(OUT)) flows through the switch 504 to an output node. A load circuit (represented here by a load capacitance and resistance in parallel) is coupled to the output node. The switching regulator 502 further includes a low-side (LS) transistor switch 514 (for example, implemented as an n-channel MOS transistor) having a drain terminal coupled, preferably directly connected, to the switching node 506 and a source terminal coupled, preferably directly connected, to a reference voltage node (for example, ground). The gate of the LS transistor switch 514 is driven by a low side driver circuit 518 that outputs a low side control signal 520. A current I_(LS) flows through the switch 514. In a preferred embodiment, the high side control signal 510 and the low side control signal 520 have a same frequency and are non-overlapping pulse width modulated (PWM) signals where the low side control signal 520 controls the LS transistor switch 514 to turn on during a switching on phase (T_(ON)) and turn off during a switching off phase (T_(OFF)). In an embodiment, the high side control signal 510 may be derived from the low side control signal 520 with a small margin (referred to as dead time) to avoid both transistors being on at the same time. The high side control signal 510 controls the HS transistor switch 504 to turn on during a portion of the switching off phase (T OFF) which is delimited by a dead time 522 due to the non-overlapping switch control. An inductor 530 has a first terminal coupled, preferably directly connected, to the switching node 506 and a second terminal coupled, preferably directly connected, to receive an input voltage Vin. A current I_(coil) (also referred to herein as an input current I_(IN)) flows through the inductor 530.

The input current detection circuit 500 includes a coil current detection circuit 540 with a low side current sensing circuit configured to sense the current I_(LS) flowing through the LS transistor switch 514 and output a signal that is indicative of the extrapolated value I_(INextrap) for the input current I_(IN) of the switching regulator.

Reference is now made to FIG. 19 which shows a circuit diagram for an embodiment of the coil current detection circuit 540. A first input transistor M31 (for example, implemented as an n-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the drain terminal (d) of the LS transistor switch 514 (at the switching node 506) and a drain terminal coupled, preferably directly connected, to a first (non-inverting (+)) input of a differential amplifier 542. The gate terminal of the first input transistor M31 is driven by the low side control signal 520. A second input transistor M32 (for example, implemented as an n-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the source terminal (s) of the LS transistor switch 514 (at the reference node; ground) and a drain terminal coupled, preferably directly connected, to a second (inverting (−)) input of the differential amplifier 542. The gate terminal of the second input transistor M32 is also driven by the low side control signal 520. Input transistors M31 and M32 are controlled by the low side control signal 520 to turn on during the switching on phase (T_(ON)). An output transistor M33 (for example, implemented as an n-channel MOS transistor) has a source terminal coupled, preferably directly connected, in a negative feedback circuit to the second (inverting) input of the differential amplifier 542. A drain terminal of the output transistor M33 is connected to an input leg of a current mirroring circuit 543 (formed by mirror connected p-channel MOS transistors). The foregoing circuitry forms a low side current sensing circuit. A resistor 544 has a first terminal coupled, preferably directly connected, to an output leg of the current mirroring circuit 543 and a second terminal coupled, preferably directly connected, to the reference voltage node (for example, ground). A switch transistor M34 (for example, implemented as an n-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the first terminal of resistor R544 (at the output of the current mirror) and a drain terminal coupled, preferably directly connected, to an output 556 of the coil current detection circuit 540 producing the signal Vsense indicative of the input current I_(IN) of the switching regulator. The gate terminal of the switch transistor M34 is driven by a signal derived from the low side control signal 520. A capacitor 552 is coupled, preferably directly connected, between the output 556 (at the drain terminal of the switch transistor M34) and the reference voltage node. The capacitor 552 is selectively charged in response to the current I_(sense) when the switch transistor M34 is turned on by the signal derived from the low side control signal 520 during the switching on phase (T_(ON)). The charging current for capacitor 552 is equal to the current I_(sense) minus the current flowing through the resistor 544.

The first and second input transistors M31, M32 function to sample the source and drain voltages, respectively, of the LS transistor switch 514 responsive to the low side control signal 520 only during switching on phase (T_(ON)) for application to the differential amplifier 542. The difference between those sampled voltages is converted by the differential amplifier 542 and output transistor M33 to a current I_(sense) which replicates the current flowing through the inductor 530 during switching on phase (T_(ON)). This current I_(sense) is converted to a voltage by resistor 544. That voltage is selectively filtered in response to actuation of the switch transistor M34 using capacitor 552 to generate the output voltage signal Vsense. Such filter action is effective when the time constant is substantially longer than the ON time: R₅₄₄ C₅₅₂>>T_(ON).

An advantage of the circuit implementation shown in FIGS. 18 and 19 is that it obviates the need for the duty cycle detect and divide circuit 180 used in the circuit implementation of FIGS. 8, 10, 11 and 12 .

FIG. 5B shows a circuit diagram for an embodiment of the differential amplifier 542.

Let K be a ratio of power to power sense. During the switching on phase (T_(ON)) when the low side control signal 520 is asserted, transistors 514, M31, and M32 are all turned on. If the gain of the differential amplifier 542 is high enough to ensure that the voltages at the inverting (−) input and non-inverting (+) input of the differential amplifier 542 are substantially equal (through the negative feedback with transistor M33), the following is true:

V ₅₀₆ −K*R _(on) *I _(bias) =−K*R _(on)*(I _(bias) −I _(sense))

Where: V 506 is the voltage at the switching node 506, R_(on) is drain-to-source resistance of the transistor 514 in the ON state (i.e., Rds_on), I_(bias) is the bias current for the current source circuits 161, 163, and I_(sense) is the current flowing through the transistor M33.

Then, the sense current I_(sense) is:

$I_{sense} = {\frac{V_{506}}{K*R_{on}} = \frac{I_{LS}}{K}}$

Where: I_(LS) is the current flowing through the inductor 530 when the transistor 514 in the ON state.

It will accordingly be noted that the current I_(LS) and its replica current I_(sense) increase with a ratio of K. In this context, K is substantially constant, and it allows to have the current I_(sense) directly proportional to the current flowing through the inductor 530 when the LS transistor switch 514 is in the ON state. It is important that the value for K is large enough to have the magnitude of I_(sense) in a range of some micro-amperes, despite an inductor current I_(LS) that has a magnitude on the order of amperes.

Due to the switching operation of the switch transistor M34, filtering is performed by capacitor 552 only during switching on phase (T_(ON)). The average of the current in the capacitor 552 is zero at steady state. Considering the capacitance of the capacitor 552 to be high enough that the voltage across the capacitor (Vsense) is substantially constant, the current flowing through the capacitor 552 during switching on phase (T_(ON)) is given by the following formula:

${I_{c}(t)} = {{I_{sense}(t)} - \frac{Vsense}{R}}$

Where: R is the resistance of the resistor 544.

Then:

${\frac{1}{T_{ON}}{\int_{0}^{T_{ON}}{{I_{sense}(t)}dt}}} = \frac{Vsense}{R}$

Solving for Vsense:

${Vsense} = {{\frac{R}{K}\frac{1}{T_{ON}}{\int_{0}^{T_{ON}}{{I_{LS}(t)}dt}}} = {\frac{R}{K}I_{IN}}}$

From the foregoing, the output voltage Vsense from circuit 540, which is indicative of the input current I_(IN), is directly proportional to the average of the current flowing through the LS transistor switch 514 responsive to assertion of the low side control signal 520 during switching on phase (T_(ON)).

Reference is now made to FIG. 20 which shows a circuit diagram for a modified embodiment of the coil current detection circuit 540 that is used in connection with DCM operation. A first input transistor M31 (for example, implemented as an n-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the drain terminal (d) of the LS transistor switch 514 (at the switching node 506) and a drain terminal coupled, preferably directly connected, to a first (non-inverting (+)) input of a differential amplifier 542. The gate terminal of the first input transistor M31 is driven by the low side control signal 520. A second input transistor M32 (for example, implemented as an n-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the source terminal (s) of the LS transistor switch 514 (at the reference node; ground) and a drain terminal coupled, preferably directly connected, to a second (inverting (−)) input of the differential amplifier 542. The gate terminal of the second input transistor M32 is also driven by the low side control signal 520. Input transistors M31 and M32 are controlled by the low side control signal 520 to turn on during the switching on phase (T_(ON)). An output transistor M33 (for example, implemented as an n-channel MOS transistor) has a source terminal coupled, preferably directly connected, in a negative feedback circuit to the second (inverting) input of the differential amplifier 542. A drain terminal of the output transistor M33 is connected to an input leg of a current mirroring circuit 543 (formed by mirror connected p-channel MOS transistors). The foregoing circuitry forms a low side current sensing circuit. A resistor 544 has a first terminal coupled, preferably directly connected, to an output leg of the current mirroring circuit 543 and a second terminal coupled, preferably directly connected, to the reference voltage node (for example, ground). A first switch transistor M34 (for example, implemented as an n-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the first terminal of the resistor R₅₄₄ at the output of the current mirror and a drain terminal coupled, preferably directly connected, to a first intermediate node 570. The gate terminal of the first switch transistor M34 is driven by a signal derived from the low side control signal 520. A capacitor 552 is coupled, preferably directly connected, between node 570 (at the drain terminal of the first switch transistor M34) and the reference voltage node. The capacitor 552 is selectively charged in response to the current Isense when the switch transistor M34 is turned on by the signal derived from the low side control signal 520 during the switching on phase (T_(ON)). The charging current for capacitor 552 is equal to the current Isense minus the current flowing through the resistor 544. A second switch transistor M35 (for example, implemented as an n-channel MOS transistor) has a source terminal coupled, preferably directly connected, to the node 570 (at the drain terminal of the first switch transistor M34) and a drain terminal coupled, preferably directly connected, to a second intermediate node 572. The gate terminal of the second switch transistor M35 is driven by a signal derived from a logical inversion of a control signal High-Z, generated by a timing control circuit, which is logic high when the regulator circuit is in the high-impedance (High-Z) state and is logic low when the regulator circuit is not in the high-impedance (High-Z) state. A third switch transistor M36 (for example, implemented as an n-channel MOS transistor) has a drain terminal coupled, preferably directly connected, to the node 572 (at the drain terminal of the second switch transistor M35) and a source terminal coupled, preferably directly connected, to the reference voltage node. The gate terminal of the third switch transistor M36 is driven by a control signal High-Z′, generated by the timing control circuit, which is derived from the signal High-Z but with a small dead time (for example, on the order of a few hundred pico-seconds in order to avoid cross-conduction with transistors M35 and M36). A resistor 574 is coupled between the node 572 and the output 556 of the coil current detection circuit 540 which produces the signal Vsense indicative of the input current I_(IN) of the switching regulator. A capacitor 576 is coupled, preferably directly connected, between the output 556 and the reference voltage node. The resistor 574 and capacitor 576 form a passive low-pass filter circuit.

The circuitry formed by the first switching transistor M34 and capacitor 552 operates, analogous to that discussed above in connection with FIG. 19 , to generate a signal at node 570 that is proportional to an average of the input current I_(LS) flowing through the low side transistor 514 during switching on phase (T_(ON)).

The circuitry formed by the second and third switching transistors M35, M36 functions as a masking circuit that masks (or blocks) the contribution of the signal at node 570 during the high impedance (High-Z) state. During the high impedance (High-Z) state, the switching transistor M35 is deactuated (blocking the signal at node 570 from passing to node 572) and the switching transistor M36 is actuated (connecting node 572 to ground). Conversely, when not in the high impedance (High-Z) state, the switching transistor M35 is actuated (passing the signal at node 570 to node 572 to be filtered by resistor 574 and capacitor 576) and the switching transistor M36 is deactuated (disconnecting node 572 from ground).

The circuitry formed by resistor 574 and capacitor 576 filters the signal at node 572 to generate the output voltage Vsense which is proportional to the input current I_(IN) of the switching regulator.

Operationally, the average of the currents flowing through capacitors 552 and 576 must be zero. Thus, the following system of equations applies:

$\left| \left\{ \begin{matrix} {{{\int_{0}^{T_{ON}}{{I_{sense}(t)}dt}} - {\frac{V_{A}}{R_{444}}T_{ON}} + {\frac{{Vsense} - V_{A}}{R_{474}}\left( {T_{ON} + T_{OFF}} \right)}} = 0} \\ {{{\frac{V_{A} - {Vsense}}{R_{474}}\left( {T_{ON} + T_{OFF}} \right)} - {\frac{Vsense}{R_{474}}\left( {T_{SW} - T_{ON} - T_{OFF}} \right)}} = 0} \end{matrix} \right. \right.$

Where: V A is the voltage at node 570; R₅₄₄ is the resistance of resistor 544; R574 is the resistance of resistor 574; T_(ON) is the on time; T_(OFF) is the off time; and T_(SW) is the overall switching period which includes T_(ON), T_(OFF) and High-Z.

It is assumed here that either the resistance R₄₇₄ is very high, or that there is voltage buffering using voltage follower 480. The following simplification then applies:

$\left\{ \begin{matrix} {{{\int_{0}^{T_{ON}}{{I_{sense}(t)}dt}} - {\frac{V_{A}}{R_{444}}T_{ON}}} = 0} \\ {{{\frac{V_{A} - {Vsense}}{R_{474}}\left( {T_{ON} + T_{OFF}} \right)} - {\frac{Vsense}{R_{474}}\left( {T_{SW} - T_{ON} - T_{OFF}} \right)}} = 0} \end{matrix} \right.$

Solving this system of equations for Vsense:

${Vsense} = {\frac{\int_{0}^{T_{ON}}{{I_{LS}(t)}{dt}}}{K\left\lbrack {\frac{1}{R_{544}}T_{SW}\frac{T_{ON}}{T_{ON} + T_{OFF}}} \right\rbrack} = {\frac{R_{544}}{K}\frac{1}{T_{SW}}\frac{T_{ON} + T_{OFF}}{T_{ON}}{\int_{0}^{T_{ON}}{{I_{LS}(t)}dt}}}}$

Where:

$I_{IN} = {\frac{T_{ON} + T_{OFF}}{T_{SW}}\frac{1}{T_{ON}}{\int_{0}^{T_{ON}}{{I_{LS}(t)}dt}}}$

Substituting:

${Vsense} = {\frac{R_{544}}{K}I_{IN}}$

Thus, the voltage of the output signal Vsense is directly proportional to the input current I_(IN) flowing in the inductor.

As noted above, the solution is based on the assumption that the resistance R574 for resistor 574 is substantially high or a unity gain buffer circuit 580 is inserted in the signal path between nodes 570 and 572 in series with the second switch transistor M35.

While the invention has been illustrated and described in detail in the drawings and foregoing description, such illustration and description are considered illustrative or exemplary and not restrictive; the invention is not limited to the disclosed embodiments. Other variations to the disclosed embodiments can be understood and effected by those skilled in the art in practicing the claimed invention, from a study of the drawings, the disclosure, and the appended claims. 

1. A switching regulator circuit, comprising: a high side (HS) transistor coupled between a first node and a switching node; a low side (LS) transistor coupled between the switching node and a second node; driver circuitry for controlling actuation of the HS transistor and LS transistor in accordance with a duty cycle including a switching on phase and a switching off phase with a first dead time delimiting the switching off phase; an inductor having a first terminal coupled to the switching node and a second terminal coupled to a third node, wherein a coil current flows through the inductor; and a coil current detection circuit comprising: a current sensing circuit configured to sense a transistor current flowing through only one of the HS transistor and LS transistor during the switching on phase and generate a current sense signal; a first capacitor; a first switch configured to selectively charge the first capacitor in response to the current sense signal only during the switching on phase.
 2. The circuit of claim 1, wherein the switching regulator circuit is a buck-type regulator circuit, wherein the first node is configured to receive an input voltage, the second node is configured to receive a reference voltage and the third node is an output voltage node, and wherein the coil current is output current from the buck-type regulator circuit.
 3. The circuit of claim 1, wherein the switching regulator circuit is a boost-type regulator circuit, wherein the first node is an output voltage node, the second node is configured to receive a reference voltage and the third node is configured to receive an input voltage, and wherein the coil current is input current to the boost-type regulator circuit.
 4. The circuit of claim 1, wherein the driver circuitry controls operation in continuous conduction mode (CCM), further comprising: a low pass filter circuit; and switching circuitry that passes a voltage at the first capacitor to an input of the low pass filter circuit.
 5. The circuit of claim 1, wherein the driver circuitry controls operation in discontinuous conduction mode (DCM) where the HS transistor and LS transistor are further controlled to switch off and provide a high impedance state at the switching node, the coil current detection circuit further comprising: a filter circuit comprising a first resistor and a second capacitor; a second switch configured to selectively pass a voltage at the first capacitor to an input of the filter circuit when the switching node is not in the high impedance state; and a third switch configured to couple the input of the filter circuit to ground when the switching node is in the high impedance state.
 6. The circuit of claim 5, wherein the coil current detection circuit further comprises a buffer circuit connected between the first capacitor and the second switch.
 7. The circuit of claim 1, wherein said current sensing circuit comprises: a differential amplifier having a first input, a second input and an output; a first input transistor coupled between a first terminal of said one of the HS transistor and LS transistor and the first input of the differential amplifier; a second input transistor coupled between a second terminal of said one of the HS transistor and LS transistor and the second input of the differential amplifier; wherein said first and second input transistors are actuated during switching on phase for the duty cycle; an output transistor coupled between the first input of the differential amplifier and a first intermediate node; and a resistor coupled between the first intermediate node and the second node; wherein said current sense signal is sourced by the output transistor.
 8. The circuit of claim 1, wherein said current sensing circuit comprises: a differential amplifier having a first input, a second input and an output; a first input transistor coupled between a first terminal of said one of the HS transistor and LS transistor and the first input of the differential amplifier; a second input transistor coupled between a second terminal of said one of the HS transistor and LS transistor and the second input of the differential amplifier; wherein said first and second input transistors are actuated during switching on phase for the duty cycle; an output transistor coupled between the first input of the differential amplifier and a first intermediate node; a current mirror having an input leg coupled to the first intermediate node and an output leg coupled to a second intermediate node; and a resistor coupled between the second intermediate node and the second node; wherein said current sense signal is sourced by the output leg of the current mirror.
 9. A circuit for detecting a coil current flowing through an inductor of a switching regulator circuit having a switching transistor coupled to the inductor and driver circuitry for controlling actuation of the switching transistor in accordance with a duty cycle including a switching on phase and a switching off phase delimited by dead time, comprising: a current sensing circuit configured to sense a transistor current flowing through the switching transistor during switching on phase of the duty cycle to generate a current sense signal indicative of the sensed transistor current; and a switched capacitor circuit actuated in response to the switching on phase to charge a capacitor dependent on the current sense signal.
 10. The circuit of claim 9, wherein the switching regulator circuit is a buck-type regulator circuit, wherein the switching transistor is a high-side transistor, and wherein the coil current is output current from the buck-type regulator circuit.
 11. The circuit of claim 9, wherein the switching regulator circuit is a boost-type regulator circuit, wherein the switching transistor is a low-side transistor, and wherein the coil current is input current to the boost-type regulator circuit.
 12. The circuit of claim 9, wherein the driver circuitry controls operation in continuous conduction mode (CCM), further comprising: a low pass filter circuit; and switching circuitry that passes a voltage at the capacitor to an input of the low pass filter.
 13. The circuit of claim 9, wherein the driver circuitry controls operation in discontinuous conduction mode (DCM) where the HS transistor and LS transistor are further controlled to switch off and provide a high impedance state at the switching node, further comprising: a low pass filter circuit; a switch configured to selectively pass a voltage at the capacitor to an input of the low pass filter when the switching node is not in the high impedance state; and a further switch configured to couple the input of the low pass filter to ground when the switching node is in the high impedance state.
 14. The circuit of claim 13, further comprising a buffer circuit connected between the switched capacitor circuit and the switch.
 15. A method for detecting a coil current flowing through an inductor of a switching regulator circuit having a switching transistor coupled to the inductor and driver circuitry for controlling actuation of the switching transistor in accordance with a duty cycle including a switching on phase and a switching off phase delimited by dead time, comprising: sensing a transistor current flowing through the switching transistor during switching on phase to generate a current sense signal indicative of the sensed transistor current; and selectively charging a capacitor of a switched capacitor circuit dependent on the current sense signal during the switching on phase.
 16. The method of claim 15, wherein the switching regulator circuit is a buck-type regulator circuit, wherein the switching transistor is a high-side transistor, and wherein the coil current is output current from the buck-type regulator circuit.
 17. The method of claim 15, wherein the switching regulator circuit is a boost-type regulator circuit, wherein the switching transistor is a low-side transistor, and wherein the coil current is input current to the boost-type regulator circuit.
 18. The method of claim 15, wherein the driver circuitry controls operation in continuous conduction mode (CCM), further comprising: passing a voltage at the capacitor to an input of a low pass filter.
 19. The method of claim 15, wherein the driver circuitry controls operation in discontinuous conduction mode (DCM) where the HS transistor and LS transistor are further controlled to switch off and provide a high impedance state at the switching node, further comprising: selectively passing a voltage at the capacitor to an input of a low pass filter when the switching node is not in the high impedance state; and coupling the input the low pass filter to ground when the switching node is in the high impedance state. 